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产品型号LM2717MTX-ADJ的概述

LM2717MTX-ADJ芯片概述 LM2717MTX-ADJ是一款由德州仪器（Texas Instruments）公司设计和生产的高性能低压差（LDO）稳压器。该芯片专为满足移动设备和其他便携式电子产品对电源的严格要求而开发，具有低功耗、高效率和高输出电流能力的特点。LM2717MTX-ADJ支持可调节输出电压范围，提供灵活性以适应不同应用。这一芯片的运行电压范围在2.5V至16V之间，能够提供高达1A的输出电流，满足大多数移动设备功耗需求。LM2717MTX-ADJ具有高达75dB的输出噪声抑制比，可以在对噪声要求较高的应用场合表现出色。其内置的过流保护和热关断功能在设备工作于极端条件时，能够有效保护芯片及其它外围电路的安全运行。 LM2717MTX-ADJ的详细参数 1. 输入电压范围：2.5V至16V 2. 输出电压：可调，通常为1.225V至15V 3. 最大输出电流：1A...

产品型号LM2717MTX-ADJ的Datasheet PDF文件预览

March 2006

LM2717-ADJ

Dual Step-Down DC/DC Converter

General Description

Features

n Adjustable buck converter with a 2.2A, 0.16Ω, internal

switch (Buck 1)

The LM2717-ADJ is composed of two PWM DC/DC buck

(step-down) converters. Both converters are used to gener-

ate an adjustable output voltage as low as 1.267V. Both also

feature low R_DSON(0.16Ω) internal switches for maximum

efficiency. Operating frequency can be adjusted anywhere

between 300kHz and 600kHz allowing the use of small

external components. External soft-start pins for each con-

verter enables the user to tailor the soft-start times to a

specific application. Each converter may also be shut down

independently with its own shutdown pin. The LM2717-ADJ

is available in a low profile 24-lead TSSOP package ensur-

ing a low profile overall solution.

n Adjustable buck converter with a 3.2A, 0.16Ω, internal

switch (Buck 2)

n Operating input voltage range of 4V to 20V

n Input undervoltage protection

n 300kHz to 600kHz pin adjustable operating frequency

n Over temperature protection

n Small 24-Lead TSSOP package

Applications

n TFT-LCD Displays

n Handheld Devices

n Portable Applications

n Laptop Computers

n Automotive Applications

Typical Application Circuit

20167901

DS201679

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Connection Diagram

Top View

20167904

24-Lead TSSOP

Ordering Information

Order Number

LM2717MT-ADJ

LM2717MTX-ADJ

LM2717MT-ADJ

LM2717MTX-ADJ

Spec

Package Type

TSSOP-24

NSC Package Drawing

MTC24

Supplied As

61 Units, Rail

MTC24

2500 Units, Tape and Reel

61 Units, Rail

NOPB

MTC24

2500 Units, Tape and Reel

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Pin Descriptions

Name

PGND

AGND

FB1

Function

Power ground. PGND and AGND pins must be connected together directly at the part.

Analog ground. PGND and AGND pins must be connected together directly at the part.

Buck 1 output voltage feedback input.

V_C1

Buck 1 compensation network connection. Connected to the output of the voltage error

amplifier.

V_BG

V_C2

Bandgap connection.

Buck 2 compensation network connection. Connected to the output of the voltage error

amplifier.

FB2

AGND

PGND

SW2

Buck 2 output voltage feedback input.

Analog ground. PGND and AGND pins must be connected together directly at the part.

Power ground. PGND and AGND pins must be connected together directly at the part.

Buck 2 power switch input. Switch connected between V_INpins and SW2 pin.

Analog power input. All V_INpins are internally connected and should be connected

together directly at the part.

V_IN

Analog power input. All V_INpins are internally connected and should be connected

together directly at the part.

CB2

SHDN2

SS2

Buck 2 converter bootstrap capacitor connection.

Shutdown pin for Buck 2 converter. Active low.

Buck 2 soft start pin.

FSLCT

Switching frequency select input. Use a resistor to set the frequency anywhere between

300kHz and 600kHz.

SS1

SHDN1

CB1

Buck 1 soft start pin.

Shutdown pin for Buck 1 converter. Active low.

Buck 1 converter bootstrap capacitor connection.

V_IN

Analog power input. All V_INpins are internally connected and should be connected

together directly at the part.

SW1

Buck 1 power switch input. Switch connected between V_INpins and SW1 pin.

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Block Diagram

20167903

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Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

Power Dissipation(Note 2)

Lead Temperature

Internally Limited

300˚C

Vapor Phase (60 sec.)

Infrared (15 sec.)

215˚C

220˚C

V_IN

−0.3V to 22V

ESD Susceptibility (Note 3)

Human Body Model

SW1 Voltage

SW2 Voltage

FB1 Voltage

2kV

−0.3V to 22V

−0.3V to 7V

Operating Conditions

FB2 Voltage

−0.3V to 7V

Operating Junction

Temperature Range

(Note 4)

V_C1Voltage

1.75V ≤ V_C1≤ 2.25V

0.965V ≤ V_C2≤ 1.565V

−0.3V to 7.5V

−0.3V to 2.1V

AGND to 5V

V_C2Voltage

−40˚C to +125˚C

−65˚C to +150˚C

4V to 20V

SHDN1 Voltage

SHDN2 Voltage

SS1 Voltage

Storage Temperature

Supply Voltage

SW1 Voltage

20V

SS2 Voltage

SW2 Voltage

20V

FSLCT Voltage

Maximum Junction Temperature

Switching Frequency

300kHz to 600kHz

150˚C

Electrical Characteristics

Specifications in standard type face are for T_J= 25˚C and those with boldface type apply over the full Operating Tempera-

ture Range (T_J= −40˚C to +125˚C). V_IN= 5V, I_L= 0A, and F_SW= 300kHz unless otherwise specified.

Min

(Note 4)

Typ

(Note 5)

Max

(Note 4)

Symbol

Parameter

Conditions

Units

I_Q

Total Quiescent Current (both Not Switching

2.7

µA

switchers)

Switching, switch open

V_SHDN= 0V

V_BG

Bandgap Voltage

1.248

1.294

1.299

0.01

1.267

%/V

1.230

%V_BG/∆V_INBandgap Voltage Line

-0.01

Regulation

0.125

1.286

1.288

1.286

1.288

V_FB1

V_FB2

Buck 1 Feedback Voltage

Buck 2 Feedback Voltage

1.236

1.214

1.236

1.214

1.258

I_CL1(Note 6) Buck 1 Switch Current Limit V_IN= 8V (Note 7)

V_IN= 12V, V_OUT= 3.3V

_CL2(Note 6) Buck 2 Switch Current Limit V_IN= 8V (Note 7)

2.2

1.65

3.2

1.4

2.6

2.0

V_IN= 12V, V_OUT= 5V

V_IN= 20V

3.05

3.5

I_B1

I_B2

Buck 1 FB Pin Bias Current

(Note 8)

400

Buck 2 FB Pin Bias Current

(Note 8)

V_IN= 20V

400

V_IN

Input Voltage Range

Buck 1 Error Amp

Transconductance

Buck 2 Error Amp

Transconductance

Buck 1 Error Amp Voltage

Gain

g_m1

∆I = 20µA

1340

1360

134

µmho

g_m2

µmho

V/V

A_V1

A_V2

Buck 2 Error Amp Voltage

Gain

136

V/V

D_MAX

Maximum Duty Cycle

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Electrical Characteristics (Continued)

Specifications in standard type face are for T_J= 25˚C and those with boldface type apply over the full Operating Tempera-

ture Range (T_J= −40˚C to +125˚C). V_IN= 5V, I_L= 0A, and F_SW= 300kHz unless otherwise specified.

Min

(Note 4)

240

Typ

(Note 5)

300

Max

(Note 4)

360

720

Symbol

F_SW

Parameter

Conditions

Units

Switching Frequency

R_F= 46.4k

R_F= 22.6k

kHz

µA

480

600

I_SHDN1

I_SHDN2

I_L1

Buck 1 Shutdown Pin Current 0V V_SHDN17.5V

−5

Buck 2 Shutdown Pin Current 0V V_SHDN27.5V

−5

µA

Buck 1 Switch Leakage

Current

V_IN= 20V

0.01

160

µA

mΩ

I_L2

Buck 2 Switch Leakage

Current

V_IN= 20V

R_DSON1

R_DSON2

Th_SHDN1

Th_SHDN2

Buck 1 Switch R_DSON(Note I_SW= 100mA

180

300

180

300

Buck 2 Switch R_DSON(Note I_SW= 100mA

Buck 1 SHDN Threshold

Output High

Output Low

Output High

Output Low

1.8

1.36

1.33

1.36

1.33

0.7

Buck 2 SHDN Threshold

0.7

I_SS1

I_SS2

UVP

Buck 1 Soft Start Pin Current

Buck 2 Soft Start Pin Current

On Threshold

µA

3.8

3.6

Off Threshold

3.3

θ_JA

Thermal Resistance

(Note 10)

TSSOP, package only

115

˚C/W

Note 1: Absolute maximum ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions for which the device is intended to

be functional, but device parameter specifications may not be guaranteed. For guaranteed specifications and test conditions, see the Electrical Characteristics.

Note 2: The maximum allowable power dissipation is a function of the maximum junction temperature, T (MAX), the junction-to-ambient thermal resistance, θ

and the ambient temperature, T . See the Electrical Characteristics table for the thermal resistance. The maximum allowable power dissipation at any ambient

temperature is calculated using: P (MAX) = (T

− T )/θ . Exceeding the maximum allowable power dissipation will cause excessive die temperature, and the

A JA

J(MAX)

regulator will go into thermal shutdown.

Note 3: The human body model is a 100 pF capacitor discharged through a 1.5kΩ resistor into each pin.

Note 4: All limits guaranteed at room temperature (standard typeface) and at temperature extremes (bold typeface). All room temperature limits are 100% tested

or guaranteed through statistical analysis. All limits at temperature extremes are guaranteed via correlation using standard Statistical Quality Control (SQC) methods.

All limits are used to calculate Average Outgoing Quality Level (AOQL).

Note 5: Typical numbers are at 25˚C and represent the most likely norm.

Note 6: Duty cycle affects current limit due to ramp generator.

Note 7: Current limit at 0% duty cycle. See TYPICAL PERFORMANCE section for Switch Current Limit vs. Input Voltage.

Note 8: Bias current flows into FB pin.

Note 9: Includes the bond wires and package leads, R

from V pin(s) to SW pin.

DSON

Note 10: Refer to National’s packaging website for more detailed thermal information and mounting techniques for the TSSOP package.

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Typical Performance Characteristics

Switching I_Qvs. Input Voltage

(F_SW= 300kHz)

Shutdown I_Qvs. Input Voltage

20167960

20167961

Switching Frequency vs. Input Voltage

(F_SW= 300kHz)

Buck 1 R_DS(ON)vs. Input Voltage

20167962

20167963

Buck 1 Efficiency vs. Load Current

(V_OUT= 3.3V)

Buck 2 R_DS(ON)vs. Input Voltage

20167965

20167964

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Typical Performance Characteristics (Continued)

Buck 2 Efficiency vs. Load Current

(V_OUT= 15V)

Buck 2 Efficiency vs. Load Current

(V_OUT= 5V)

20167966

20167967

Buck 1 Switch Current Limit vs. Input Voltage

Buck 2 Switch Current Limit vs. Input Voltage

20167969

20167968

Buck 1 Switch Current Limit vs. Temperature

(V_IN= 12V)

Buck 2 Switch Current Limit vs. Temperature

(V_IN= 12V)

20167912

20167913

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Typical Performance Characteristics (Continued)

Buck 1 Switch ON Resistance vs. Temperature

Buck 2 Switch ON Resistance vs. Temperature

20167910

20167911

Switching Frequency vs. R_FResistance

20167914

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SETTING THE OUTPUT VOLTAGE

Buck Operation

The output voltage is set using the feedback pin and a

resistor divider connected to the output as shown in Figure 4.

The feedback pin voltage (V_FB) is 1.258V, so the ratio of the

feedback resistors sets the output voltage according to the

following equation:

PROTECTION (BOTH REGULATORS)

The LM2717-ADJ has dedicated protection circuitry running

during normal operation to protect the IC. The Thermal Shut-

down circuitry turns off the power devices when the die

temperature reaches excessive levels. The UVP comparator

protects the power devices during supply power startup and

shutdown to prevent operation at voltages less than the

minimum input voltage. The OVP comparator is used to

prevent the output voltage from rising at no loads allowing

full PWM operation over all load conditions. The LM2717-

ADJ also features a shutdown mode for each converter

decreasing the supply current to approximately 10µA (both in

shutdown mode).

INPUT CAPACITOR

A low ESR aluminum, tantalum, or ceramic capacitor is

needed between the input pin and power ground. This ca-

pacitor prevents large voltage transients from appearing at

the input. The capacitor is selected based on the RMS

current and voltage requirements. The RMS current is given

by:

CONTINUOUS CONDUCTION MODE

The LM2717-ADJ contains current-mode, PWM buck regu-

lators. A buck regulator steps the input voltage down to a

lower output voltage. In continuous conduction mode (when

the inductor current never reaches zero at steady state), the

buck regulator operates in two cycles. The power switch is

connected between V_INand SW1 and SW2.

In the first cycle of operation the transistor is closed and the

diode is reverse biased. Energy is collected in the inductor

and the load current is supplied by C_OUTand the rising

current through the inductor.

The RMS current reaches its maximum (I_OUT/2) when

V_INequals 2V_OUT. This value should be calculated for both

regulators and added to give a total RMS current rating. For

an aluminum or ceramic capacitor, the voltage rating should

be at least 25% higher than the maximum input voltage. If a

tantalum capacitor is used, the voltage rating required is

about twice the maximum input voltage. The tantalum ca-

pacitor should be surge current tested by the manufacturer

to prevent being shorted by the inrush current. The minimum

capacitor value should be 47µF for lower output load current

applications and less dynamic (quickly changing) load con-

ditions. For higher output current applications or dynamic

load conditions a 68µF to 100µF low ESR capacitor is rec-

ommended. It is also recommended to put a small ceramic

capacitor (0.1µF to 4.7µF) between the input pins and

ground to reduce high frequency spikes.

During the second cycle the transistor is open and the diode

is forward biased due to the fact that the inductor current

cannot instantaneously change direction. The energy stored

in the inductor is transferred to the load and output capacitor.

The ratio of these two cycles determines the output voltage.

The output voltage is defined approximately as:

where D is the duty cycle of the switch, D and D' will be

required for design calculations.

The LM2717-ADJ has a minimum switch ON time which

corresponds to a minimum duty cycle of approximately 10%

at 600kHz operation and approximately 5% at 300kHz op-

eration. In the case of some high voltage differential appli-

cations (low duty cycle operation) this minimum duty cycle

may be exceeded causing the feedback pin over-voltage

protection to trip as the output voltage rises. This will put the

device into a PFM type operation which can cause an un-

predictable frequency spectrum and may cause the average

output voltage to rise slightly. If this is a concern the switch-

ing frequency may be lowered and/or a pre-load added to

the output to keep the device full PWM operation. Note that

the OVP function monitors the FB pin so it will not function if

the feedback resistor is disconnected from the output. Due to

slight differences between the two converters it is recom-

mended that Buck 1 be used for the lower of the two output

voltages for best operation.

INDUCTOR SELECTION

The most critical parameter for the inductor in a current

mode switcher is the minimum value required for stable

operation. To prevent subharmonic oscillations and achieve

good phase margin a target minimum value for the inductor

is:

Where V_INis the minimum input voltage and R_DSONis the

maximum switch ON resistance. For best stability the induc-

tor should be in the range of 0.5L_MIN(absolute minimum)

and 2L_MIN. Using an inductor with a value less than 0.5L_MIN

can cause subharmonic oscillations. The inductor should

meet this minimum requirement at the peak inductor current

expected for the application regardless of what the inductor

ripple current and output ripple voltage requirements are. A

value larger than 2L_MINis acceptable if the ripple require-

ments of the application require it but it may reduce the

phase margin and increase the difficulty in compensating the

circuit.

DESIGN PROCEDURE

This section presents guidelines for selecting external com-

ponents.

The most important parameters for the inductor from an

applications standpoint are the inductance, peak current and

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the current limit starts at a low value and increases to

nominal at the set soft-start time. Under maximum load

conditions the output voltage may rise at the same rate as

the soft-start, however at light or no load conditions the

output voltage will rise much faster as the switch will not

need to conduct much current to charge the output capacitor.

Buck Operation (Continued)

the DC resistance. The inductance is related to the peak-to-

peak inductor ripple current, the input and the output volt-

ages (for 300kHz operation):

SHUTDOWN OPERATION (BOTH REGULATORS)

The shutdown pins of the LM2717-ADJ are designed so that

they may be controlled using 1.8V or higher logic signals. If

the shutdown function is not to be used the pin may be left

open. The maximum voltage to the shutdown pin should not

exceed 7.5V. If the use of a higher voltage is desired due to

system or other constraints it may be used, however a 100k

or larger resistor is recommended between the applied volt-

age and the shutdown pin to protect the device.

A higher value of ripple current reduces inductance, but

increases the conductance loss, core loss, and current

stress for the inductor and switch devices. It also requires a

bigger output capacitor for the same output voltage ripple

requirement. A reasonable value is setting the ripple current

to be 30% of the DC output current. Since the ripple current

increases with the input voltage, the maximum input voltage

is always used to determine the inductance. The DC resis-

tance of the inductor is a key parameter for the efficiency.

Lower DC resistance is available with a bigger winding area.

A good tradeoff between the efficiency and the core size is

letting the inductor copper loss equal 2% of the output

power.

SCHOTTKY DIODE

The breakdown voltage rating of D₁and D₂is preferred to be

25% higher than the maximum input voltage. The current

rating for the diode should be equal to the maximum output

current for best reliability in most applications. In cases

where the input voltage is much greater than the output

voltage the average diode current is lower. In this case it is

possible to use a diode with a lower average current rating,

approximately (1-D)*I_OUThowever the peak current rating

should be higher than the maximum load current.

OUTPUT CAPACITOR

The selection of C_OUTis driven by the maximum allowable

output voltage ripple. The output ripple in the constant fre-

quency, PWM mode is approximated by:

LOOP COMPENSATION

The general purpose of loop compensation is to meet static

and dynamic performance requirements while maintaining

stability. Loop gain is what is usually checked to determine

small-signal performance. Loop gain is equal to the product

of control-output transfer function and the output-control

transfer function (the compensation network transfer func-

tion). The DC loop gain of the LM2717 is usually around

55dB to 60dB when loaded. Generally speaking it is a good

idea to have a loop gain slope that is -20dB /decade from a

very low frequency to well beyond the crossover frequency.

The crossover frequency should not exceed one-fifth of the

switching frequency, i.e. 60kHz in the case of 300kHz

switching frequency. The higher the bandwidth is, the faster

the load transient response speed will potentially be. How-

ever, if the duty cycle saturates during a load transient,

further increasing the small signal bandwidth will not help.

Since the control-output transfer function usually has very

limited low frequency gain, it is a good idea to place a pole in

the compensation at zero frequency, so that the low fre-

quency gain will be relatively large. A large DC gain means

high DC regulation accuracy (i.e. DC voltage changes little

with load or line variations). The rest of the compensation

scheme depends highly on the shape of the control-output

plot.

The ESR term usually plays the dominant role in determining

the voltage ripple. Low ESR ceramic, aluminum electrolytic,

or tantalum capacitors (such as MuRata MLCC, Taiyo Yuden

MLCC, Nichicon PL series, Sanyo OS-CON, Sprague 593D,

594D, AVX TPS, and CDE polymer aluminum) is recom-

mended. An aluminum electrolytic capacitor is not recom-

mended for temperatures below −25˚C since its ESR rises

dramatically at cold temperatures. Ceramic or tantalum ca-

pacitors have much better ESR specifications at cold tem-

perature and is preferred for low temperature applications.

BOOTSTRAP CAPACITOR

A 4.7nF ceramic capacitor or larger is recommended for the

bootstrap capacitor. For applications where the input voltage

is less than twice the output voltage a larger capacitor is

recommended, generally 0.1µF to 1µF to ensure plenty of

gate drive for the internal switches and a consistently low

R_DSON

SOFT-START CAPACITOR (BOTH REGULATORS)

The LM2717-ADJ contains circuitry that can be used to limit

the inrush current on start-up of the DC/DC switching regu-

lators. This inrush current limiting circuitry serves as a soft-

start. The external SS pins are used to tailor the soft-start for

a specific application. A current (I_SS) charges the external

soft-start capacitor, C_SS. The soft-start time can be esti-

mated as:

T_SS= C_SS*0.6V/I_SS

When programming the soft-start time use the equation

given in the Soft-Start Capacitor section above. The soft-

start function is used simply to limit inrush current to the

device that could stress the input voltage supply. The soft-

start time described above is the time it takes for the current

limit to ramp to maximum value. When this function is used

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Example: V_o= 5V, R_e= 20mΩ, C_o= 100µF, R_omax= 5V/

100mA = 50Ω, R_omin= 5V/1A = 5Ω, L = 10µH, f = 300kHz:

Buck Operation (Continued)

20167916

FIGURE 1. Control-Output Transfer Function

Once the fp range is determined, R_c1should be calculated

using:

As shown in Figure 1, the example control-output transfer

function consists of one pole (fp), one zero (fz), and a double

pole at fn (half the switching frequency). The following can

be done to create a -20dB /decade roll-off of the loop gain:

Place the first pole at 0Hz, the first zero at fp, the second

pole at fz, and the second zero at fn. The resulting output-

control transfer function is shown in Figure 2.

Where B is the desired gain in V/V at fp (fz1), gm is the

transconductance of the error amplifier, and R1 and R2 are

the feedback resistors as shown in Figure 3. A gain value

around 10dB (3.3v/v) is generally a good starting point.

Example: B = 3.3 v/v, gm=1350µmho, R1 = 20 KΩ, R2 = 59

KΩ:

Bandwidth will vary proportional to the value of Rc1. Next,

Cc1 can be determined with the following equation:

20167917

FIGURE 2. Output-Control Transfer Function

The control-output corner frequencies, and thus the desired

compensation corner frequencies, can be determined ap-

proximately by the following equations:

Example: fpmin = 297 Hz, Rc1 = 20 KΩ:

The value of C_c1should be within the range determined by

fpmin/max. A higher value will generally provide a more

stable loop, but too high a value will slow the transient

response time.

The compensation network (Figure 3) will also introduce a

low frequency pole which will be close to 0Hz.

Where C_ois the output capacitance, R_eis the output capaci-

tance ESR, R_ois the load resistance, L is the inductor value,

and f is the switching frequency used.

A second pole should also be placed at fz. This pole can be

created with a single capacitor Cc2 and a shorted Rc2 (see

Figure 3). The minimum value for this capacitor can be

calculated by:

Since fp is determined by the output network, it will shift with

loading (Ro) and duty cycle. First determine the range of

frequencies (fpmin/max) of the pole across the expected

load range, then place the first compensation zero within that

range.

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in some cases may need to be adjusted some for optimum

stability or the values may need to be adjusted depending on

a particular applications bandwidth requirements.

Buck Operation (Continued)

Cc2 may not be necessary, however it does create a more

stable control loop. This is especially important with high

load currents.

LAYOUT CONSIDERATIONS

The LM2717-ADJ uses two separate ground connections,

PGND for the drivers and boost NMOS power device and

AGND for the sensitive analog control circuitry. The AGND

and PGND pins should be tied directly together at the pack-

age. The feedback and compensation networks should be

connected directly to a dedicated analog ground plane and

this ground plane must connect to the AGND pin. If no

analog ground plane is available then the ground connec-

tions of the feedback and compensation networks must tie

directly to the AGND pin. Connecting these networks to the

PGND can inject noise into the system and effect perfor-

mance.

Example: fz = 80 kHz, Rc1 = 20 KΩ:

A second zero can also be added with a resistor in series

with Cc2. If used, this zero should be placed at fn, where the

control to output gain rolls off at -40dB/dec. Generally, fn will

be well below the 0dB level and thus will have little effect on

stability. Rc2 can be calculated with the following equation:

The input bypass capacitor C_IN, as shown in Figure 4, must

be placed close to the IC. This will reduce copper trace

resistance which effects input voltage ripple of the IC. For

additional input voltage filtering, a 0.1µF to 4.7µF bypass

capacitors can be placed in parallel with C_IN, close to the V_IN

pins to shunt any high frequency noise to ground. The output

capacitors, C_OUT1and C_OUT2, should also be placed close to

the IC. Any copper trace connections for the C_OUTXcapaci-

tors can increase the series resistance, which directly effects

output voltage ripple. The feedback network, resistors

R_FB1(3)and R_FB2(4), should be kept close to the FB pin, and

away from the inductor to minimize copper trace connections

that can inject noise into the system. Trace connections

made to the inductors and schottky diodes should be mini-

mized to reduce power dissipation and increase overall effi-

ciency. For more detail on switching power supply layout

considerations see Application Note AN-1149: Layout Guide-

lines for Switching Power Supplies.

20167930

FIGURE 3. Compensation Network

Note that the values calculated here give a good baseline for

stability and will work well with most applications. The values

Application Information

Some Recommended Inductors (Others May Be Used)

Manufacturer

Inductor

Contact Information

www.coilcraft.com

800-3222645

Coilcraft

DO3316 and DT3316 series

SLF10145 series

TDK

www.component.tdk.com

847-803-6100

Pulse

P0751 and P0762 series

www.pulseeng.com

www.sumida.com

Sumida

CDRH8D28 and CDRH8D43 series

Some Recommended Input And Output Capacitors (Others May Be Used)

Manufacturer

Vishay Sprague

Taiyo Yuden

Capacitor

Contact Information

293D, 592D, and 595D series tantalum

High capacitance MLCC ceramic

ESRD seriec Polymer Aluminum Electrolytic

SPV and AFK series V-chip series

High capacitance MLCC ceramic

www.vishay.com

www.t-yuden.com

Cornell Dubilier

MuRata

www.cde.com

www.murata.com

www.national.com

Application Information (Continued)

20167958

FIGURE 4. 15V, 3.3V Output Application

20167959

FIGURE 5. 5V, 3.3V Output Application

www.national.com

Application Information (Continued)

20167915

FIGURE 6. 3.3V, 1.8V Output Application

www.national.com

Physical Dimensions inches (millimeters) unless otherwise noted

TSSOP-24 Pin Package (MTC)

For Ordering, Refer to Ordering Information Table

NS Package Number MTC24

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves

the right at any time without notice to change said circuitry and specifications.

For the most current product information visit us at www.national.com.

LIFE SUPPORT POLICY

NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS

WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL COUNSEL OF NATIONAL SEMICONDUCTOR

CORPORATION. As used herein:

1. Life support devices or systems are devices or systems

which, (a) are intended for surgical implant into the body, or

(b) support or sustain life, and whose failure to perform when

properly used in accordance with instructions for use

provided in the labeling, can be reasonably expected to result

in a significant injury to the user.

2. A critical component is any component of a life support

device or system whose failure to perform can be reasonably

expected to cause the failure of the life support device or

system, or to affect its safety or effectiveness.

BANNED SUBSTANCE COMPLIANCE

National Semiconductor manufactures products and uses packing materials that meet the provisions of the Customer Products

Stewardship Specification (CSP-9-111C2) and the Banned Substances and Materials of Interest Specification (CSP-9-111S2) and contain

no ‘‘Banned Substances’’ as defined in CSP-9-111S2.

Leadfree products are RoHS compliant.

National Semiconductor

Americas Customer

Support Center

National Semiconductor

Europe Customer Support Center

Fax: +49 (0) 180-530 85 86

National Semiconductor

Asia Pacific Customer

Support Center

National Semiconductor

Japan Customer Support Center

Fax: 81-3-5639-7507

Email: new.feedback@nsc.com

Tel: 1-800-272-9959

Email: europe.support@nsc.com

Deutsch Tel: +49 (0) 69 9508 6208

English Tel: +44 (0) 870 24 0 2171

Français Tel: +33 (0) 1 41 91 8790

Email: ap.support@nsc.com

Email: jpn.feedback@nsc.com

Tel: 81-3-5639-7560

www.national.com

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LM2717MTX-ADJ产品参数

型号:	LM2717MTX-ADJ
是否Rohs认证：	不符合
生命周期：	Transferred
包装说明：	TSSOP-24
Reach Compliance Code：	not_compliant
ECCN代码：	EAR99
HTS代码：	8542.39.00.01
风险等级：	5.17
模拟集成电路 - 其他类型：	DUAL SWITCHING CONTROLLER
控制模式：	CURRENT-MODE
控制技术：	PULSE WIDTH MODULATION
最大输入电压：	20 V
最小输入电压：	4 V
标称输入电压：	5 V
JESD-30 代码：	R-PDSO-G24
JESD-609代码：	e0
长度：	7.8 mm
湿度敏感等级：	1
功能数量：	1
端子数量：	24
最高工作温度：	125 °C
最低工作温度：	-40 °C
最大输出电流：	3.5 A
封装主体材料：	PLASTIC/EPOXY
封装代码：	SSOP
封装等效代码：	TSSOP24,.25
封装形状：	RECTANGULAR
封装形式：	SMALL OUTLINE, SHRINK PITCH
峰值回流温度（摄氏度）：	260
认证状态：	Not Qualified
座面最大高度：	1.1 mm
子类别：	Switching Regulator or Controllers
表面贴装：	YES
切换器配置：	SINGLE
最大切换频率：	720 kHz
温度等级：	AUTOMOTIVE
端子面层：	Tin/Lead (Sn/Pb)
端子形式：	GULL WING
端子节距：	0.65 mm
端子位置：	DUAL
处于峰值回流温度下的最长时间：	40
宽度：	4.4 mm
Base Number Matches：	1

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