LM3917芯片引脚定义引脚图及功能IC贸易商引脚图技术参数-中国IC网-芯三七

December 1994

LM391 Audio Power Driver

General Description

The LM391 audio power driver is designed to drive external

power transistors in 10 to 100 watt power amplifier designs.

High power supply voltage operation and true high fidelity

performance distinguish this IC. The LM391 is internally pro-

tected for output faults and thermal overloads; circuitry pro-

viding output transistor protection is user programmable.

Features

Y

g

High Supply Voltage

50V max

0.01%

3 mV

Y

Low Distortion

Y

Low Input Noise

Y

High Supply Rejection

90 dB

Y

Gain and Bandwidth Selectable

Y

Dual Slope SOA Protection

Y

Shutdown Pin

Equivalent Schematic and Connection Diagram

TL/H/7146–1

Dual-In-Line Package

TL/H/7146–2

Top View

Order Number LM391N-100

See NS Package Number N16A

C

1995 National Semiconductor Corporation

TL/H/7146

RRD-B30M115/Printed in U. S. A.

Absolute Maximum Ratings

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales

Office/Distributors for availability and specifications.

Package Dissipation (Note 1)

Storage Temperature

1.39W

b

a

65 C to 150 C

§

0 C to 70 C

§

a

Operating Temperature

Lead Temp. (Soldering, 10 sec.)

Thermal Resistance

§

Supply Voltage

LM391N-100

260 C

§

a

50V or 100V

g

Input Voltage

Supply Voltage less 5V

1 mA

i

20 C/W

§

JC

Shutdown Current (Pin 14)

i

63 C/W

§

JA

a

b

and V^b

90% V

e

.)

MAX

e

Electrical Characteristics T

25 C (The following are for V

§

90% V

A

MAX

Parameter

Conditions

Min

Typ

Max

Units

Quiescent Current

LM391N-100

Current in Pin 15

e

mA

V

0

5

6

IN

V^a

V^b

7

V^a

V^b

5

V

b

a

b

a

Output Swing

Positive

Negative

5

Drive Current

Source (Pin 8)

Sink (Pin 5)

5

mA

5

Noise (20 Hz–20 kHz)

Supply Rejection

Input Referred

3

mV

70

90

dB

e

Total Harmonic Distortion

f

1 kHz

20 kHz

0.01

0.10

%

0.25

Intermodulation Distortion

Open Loop Gain

60 Hz, 7 kHz, 4:1

0.01

5500

0.1

5

%

V/V

mA

mV

mA

e

f

1 kHz

1000

Input Bias Current

1.0

20

Input Offset Voltage

Positive Current Limit V

BE

Pin 10–9

Pin 9–13

Pin 10

650

10

Negative Current Limit V

BE

Positive Current Limit Bias Current

Negative Current Limit Bias Current

Pin 14 Current Comments

100

Pin 13

10

Minimum pin 14 current required for shutdown is 0.5 mA, and must not exceed 1 mA.

Maximum pin 14 current for amplifier not shut down is 0.05 mA.

The typical shutdown switch point current is 0.2 mA.

Note 1: For operation in ambient temperatures above 25 C, the device must be derated based on a 150 C maximum junction temperature and a thermal resistance

§

of 90 C/W junction to ambient.

§

Typical Applications

TL/H/7146–3

FIGURE 1. LM391 with External ComponentsÐProtection Circuitry Not Shown

2

Typical Performance Characteristics

Total Harmonic Distortion vs

e

Total Harmonic Distortion vs

e

4X)

Output Power vs Supply Voltage

Frequency (R

8X)

Frequency (R

L

Input Referred Power Supply

Rejection vs Frequency

Total Harmonic Distortion vs

AB Bias Current

Open Loop Gain vs Frequency

TL/H/7146–4

Pin Descriptions

Pin No.

Pin Name

Comments

a

b

1

2

Input

Audio input

Feedback input

Sets the dominant pole

3

Compensation

Ripple Filter

Sink Output

BIAS

4

Improves negative supply rejection

Drives output devices and is emitter of AB bias V multiplier

5

BE

6

Base of V multiplier

BE

7

BIAS

Collector of V multiplier

BE

8

Source Output

Output Sense

Drives output devices

9

Biases the IC and is used in protection circuits

Base of positive side protection circuit transistor

Diode used for dual slope SOA protection

Base of negative side protection circuit transistor

Shuts off amplifier when current is pulled out of pin

Positive supply

a

b

10

11

12

13

14

15

16

Current Limit

SOA Diode

Current Limit

Shutdown

V^a

V^b

Negative supply

3

External Components (Figure 1)

Component

Typical Value

Comments

C

IN

1 mF

Input coupling capacitor sets a low frequency pole with R

.

IN

1

e

f

L

2qR

C

IN IN

Sets input impedance and DC bias to input.

Feedback resistor; for minimum offset voltage at the output this should be equal to R

R

100k

5.1k

IN

.

IN

f

2

Feedback resistor that works with R to set the voltage gain.

f

2

1

R

f

2

1

e

a

1

A

V

C

f

10 mF

Feedback capacitor. This reduces the gain to unity at DC for minimum offset voltage at the

.

output. Also sets a low frequency pole with R

f

1

e

f

L

2qR

C

f

1

C

5 pF

Compensation capacitor. Sets gain bandwidth product and a high frequency pole.

1

GBW

e

&

GBW

, f

h

2q5000C

Max f for stable design

h

A

V

500 kHz.

C

R

C

3.9k

AB bias resistor.

A

10k

AB bias potentiometer. Adjust to set bias current in the output stage.

B

0.1 mF

5 pF

Bypass capacitor for bias. This improves high frequency distortion and transient response.

AB

R

Ripple capacitor. This improves negative supply rejection at midband and high frequencies.

, if used, must equal C .

C

R

100X

2.7X

Bleed resistor. This removes stored charge in output transistors.

eb

Output compensation resistor. This resistor and C compensate the output stage. This value

O

will vary slightly for different output devices.

O

C

R

0.1 mF

0.3X

Output compensation capacitor. This works with R to form a zero that cancels f of the

O b

output power transistors.

O

Emitter degeneration resistor. This resistor gives thermal stability to the output stage

quiescent current. IRC PW5 type.

E

39k

Shutdown resistor. Sets the amount of current pulled out of pin 14 during shutdown.

Compensation capacitors for protection circuitry.

TH

C , C

2

1000 pF

Ê₂

X

L

10X 5 mH

Used to isolate capacitive loads, usually 20 turns of wire wrapped around a 10X, 2W resistor.

ll

4

Application Hints

To prevent thermal runaway of the AB bias current the fol-

lowing equation must be valid:

GENERALIZED AUDIO POWER AMP DESIGN

Givens: Power Output

Load Impedance

a

R

E

(b

1)

MIN

s

i

(5)

JA

V

(K)

CEQMAX

Input Sensitivity

where:

is the thermal resistance of the driver transistor, junc-

Input Impedance

i

JA

tion to ambient, in C/W.

Bandwidth

§

is the emitter degeneration resistance in ohms.

The power output and load impedance determine the power

supply requirements. Output signal swing and current are

found from:

R

E

b

is that of the output transistor.

min

V

CEQMAX

equation (3).

is the highest possible value of one supply from

e

V

2 R P

L O

0

Opeak

(1)

(2)

K is the temperature coefficient of the driver base-emitter

voltage, typically 2 mV/ C.

2 P

R

O

e

§

Often the value of R is to be determined and equation (5)

is rearranged to be:

I

Opeak

0

L

E

Add 5 volts to the peak output swing (V ) for transistor

OP

a

5V) at a current

. The regulation of the supply determines the unload-

g

voltage to get the supplies, i.e.,

of I

(V

OP

i

(V ) K

JA CEQMAX

t

R

(6)

peak

E

a

b

1

MIN

ed voltage, usually about 15% higher. Supply voltage will

also rise 10% during high line conditions.

The maximum average power dissipation in each output

transistor is:

&

a

5) (1 regulation) (1.1) (3)

g

max supplies

(V

Opeak

e

P

0.4 P

OMAX

(7)

DMAX

The input sensitivity and output power specs determine the

required gain.

The power dissipation in the driver transistor is:

P

R

V

0

DMAX

O

L

ORMS

e

t

e

P

(8)

A

(4)

DRIVER(MAX)

V

b

V

MIN

IN

INRMS

Heat sink requirements are found using the following formu-

las:

Normally the gain is set between 20 and 200; for a 25 watt,

8 ohm amplifier this results in a sensitivity of 710 mV and 71

mV, respectively. The higher the gain, the higher the THD,

as can be seen from the characteristics curves. Higher gain

also results in more hum and noise at the output.

b

T

AMAX

JMAX

s

i

(9)

JA

P

D

b

i

CS

i

(10)

SA

JA

JC

The desired input impedance is set by R . Very high values

IN

can cause board layout problems and DC offsets at the out-

where:

T

is the maximum transistor junction temperature.

is the maximum ambient temperature.

put. The bandwidth requirements determine the size of C

f

and C as indicated in the external component listing.

C

jMAX

T

i

AMAX

The output transistors and drivers must have a breakdown

voltage greater than the voltage determined by equation (3).

The current gain of the drive and output device must be high

is thermal resistance junction to ambient.

is thermal resistance sink to ambient.

is thermal resistance junction to case.

JA

SA

JC

CS

enough to supply I

with 5 mA of drive from the LM391.

Opeak

is thermal resistance case to sink, typically 1 C/W for

§

The power transistors must be able to dissipate approxi-

mately 40% of the maximum output power; the drivers must

dissipate this amount divided by the current gain of the out-

puts. See the output transistor selection guide, Table A.

most mountings.

5

Application Hints (Continued)

PROTECTION CIRCUITRY

resistor is set to limit the current to less than 1 mA (the

absolute maximum). This resistor with the capacitor gives a

time constant of RC. The turn-ON delay is approximately 2

time constants.

The protection circuits of the LM391 are very flexible and

should be tailored to the output transistor’s safe operating

area. The protection V-I characteristics, circuitry, and resis-

tor formulas are described below. The diodes from the out-

put to each supply prevent the output voltage from exceed-

ing the supplies and harming the output transistors. The out-

put will do this if the protection circuitry is activated while

driving an inductive load.

Example:

Amplifier with maximum supply of 30V, like the 20W, 8X

example in the data sheet, requiring a delay of 1 second.

e

Time delay

2 RC

Max V^a

1 mA

e

R

TURN-ON DELAY

It is often desirable to delay the turn-ON of the power ampli-

fier. This is easily implemented by putting a resistor in series

with a capacitor from pin 14 to ground. The value of the

So:

e

R

a 30V rating.

e

30k. Solving for C gives 16.7 mF. Use C

20 mF with

Protection Circuitry with External Components

Protection Characteristics

TL/H/7146–6

TL/H/7146–5

a

e

Protection Circuit Resistor Formulas (V

V

)

B

Type of Protection

R , R

E

R , R

Ê₁

1

R , R

Ê₂

2

R , R

3

Ê

Ê₃

w

e

Current Limit

R

Not Required

Short

Not Required

V^a

E

I

L

b

w

V

w

Single Slope SOA

Protection

M

e

R

1 kX

1

2

I

w

#

J

L

Dual Slope SOA

b

w

V

e

b

1

Protection

a

R

E

R

1 kX

R

3

R

2

1

Ê

b

w

I

w

I

_LR

Ð

(

L

E

e

(V

V

)

B

Note: w is the current limit V voltage, 650 mV. Assumptions: V^a

transistors.

w, V

w. V is the load supply voltage. V is the maximum rated V of the output

a

ll

BE

M

CE

6

OSCILLATIONS & GROUNDING

Application Hints (Continued)

Most power amplifiers work the first time they are turned on.

They also tend to oscillate and have excess THD. Most os-

cillation problems are due to inadequate supply bypassing

and/or ground loops. A 10 mF, 50V electrolytic on each

power supply will stop supply-related oscillations. However,

if the signal ground is used for these bypass caps the THD

is usually excessive. The signal ground must return to the

power supply alone, as must the output load ground. All

other groundsÐbypass, output R-C, protection, etc., can tie

together and then return to supply. This ground is called

high frequency ground. On the 40W amplifier schematic all

the grounds are labeled.

TRANSIENT INTERMODULATION DISTORTION

There has been a lot of interest in recent years about tran-

sient intermodulation distortion. Matti Otala of University of

Oulu, Oulu, Finland has published several papers on the

subject. The results of these investigations show that the

open loop pole of the power amplifier should be above 20

kHz.

To do this with the LM391 is easy. Put a 1 MX resistor from

pin 3 to the output and the open loop gain is reduced to

about 46 dB. Now the open loop pole is at 30 kHz. The

current in this resistor causes an offset in the input stage

that can be cancelled with a resistor from pin 4 to ground.

The resistor from pin 4 to ground should be 910 kX rather

than 1 MX to insure that the shutdown circuitry will operate

correctly. The slight difference in resistors results in about

15 mV of offset. The 40W, 8X amplifier schematic shows

the hookup of these two resistors.

Capacitive loads can cause instabilities, so they are isolated

from the amplifier with an inductor and resistor in the output

lead.

AB BIAS CURRENT

To reduce distortion in the output stage, all the transistors

are biased ON slightly. This results in class AB operation

and reduces the crossover (notch) distortion of the class B

stage to a low level, (see performance curve, THD vs AB

BRIDGE AMPLIFIER

A switch can be added to convert a stereo amplifer to a

single bridge amplifer. The diagram below shows where the

switch and one resistor are added. When operating in the

bridge mode the output load is connected between the two

bias). The potentiometer, R , from pins 6–7 is adjusted to

B

give about 25 mA of current in the output stage. This current

is usually monitored at the supply or by measuring the volt-

Ý

2 is disconnected.

outputs, the input is V

1, and V

age across R .

E

IN

Typical Applications (Continued)

Bridge Circuit Diagram

TL/H/7146–7

Output Transistors Selection Guide

Table A.

Driver Transistor

Output Transistor

Power

Output

PNP

NPN

PNP

NPN

@

20W 8X

MJE711

MJE171

D43C8

MJE721

MJE181

D42C8

TIP42A

2N6490

TIP41A

2N6487

@

30W 4X

@

40W 8X

MJE712

MJE172

D43C11

MJE722

MJE182

D42C11

2N5882

2N5880

@

60W 4X

7

Application Hints (Continued)

A 20W, 8X; 30W, 4X AMPLIFIER

Givens:

Solving for C :

f

1

t

e

C

f

7.8 mF; use 10 mF

Power Output

20W into 8X

30W into 4X

2qR f

f L

1

The recommended value for C is 5 pF for gains of 20 or

C

Input Sensitivity

Input Impedance

1V Max

100k

larger. This gives a gain-bandwidth product of 6.4 MHz and

a resulting bandwidth of 320 kHz, better than required.

The breakdown voltage requirement is set by the maximum

supply; we need a minimum of 58V and will use 60V. We

must now select a 60V power transistor with reasonable

g

20 Hz–20 kHz 0.25 dB

Bandwidth

Equations (1) and (2) give:

e

20W/8X

30W/4X

V

17.9V

15.5V

I

2.24A

3.87A

OP

beta at I

, 3.87A. The TIP42, TIP41 complementary pair

Opeak

are 60V, 60W transistors with a minimum beta of 30 at 4A.

The driver transistor must supply the base drive given 5 mA

drive from the LM391. The MJE711, MJE721 complementa-

ry driver transistors are 60V devices with a minimum beta of

40 at 200 mA. The driver transistors should be much faster

OP

Therefore the supply required is:

@

23V 2.24A, reducing to . . .

g

@

21V 3.87A

g

With 15% regulation and high line we get 29V from equa-

tion (3).

(higher f ) than the output transistors to insure that the R-C

T

on the output will prevent instability.

Sensitivity and equation (4) set minimum gain:

To find the heat sink required for each output transistor we

use equations (7), (9), and (10):

c

20

8

0

t

e

A

12.65

V

e

P

0.4 (30)

12W

(7)

1

D

b

150 C 55 C

§

s

e

i

7.9 C/W for T

55 C (9)

§

We will use a gain of 20 with resulting sensitivity of 632 mV.

§

JA

AMAX

e

7.9 2.1 1.0 4.8 C/W

12

Letting R equal 100k gives the required input impedance.

IN

For low DC offsets at the output we let R

for R gives:

s

b

i

(10)

§

SA

e

100k. Solving

f

2

If both transistors are mounted on one heat sink the thermal

resistance should be halved to 2.4 C/W.

f

1

§

The maximum average power dissipation in each driver is

found using equation (8):

e

R

100k

5.26k; use 5.1k

f

2

100k

e

R

f

1

b

20

1

12

e

400 mW

P

The bandwidth requirement must be stated as a pole, i.e.,

the 3 dB frequency. Five times away from a pole gives 0.17

dB down, which is better than the required 0.25 dB. There-

fore:

DRIVER(MAX)

30

Using equation (9):

b

155 55

s

e

i

237 C/W

§

JA

20

0.4

e

4 Hz

f

L

5

e

c

e

5 100 kHz

f

20k

h

8

Application Hints (Continued)

Since the free air thermal resistance of the MJE711,

MJE721 is 100 C/W, no heat sink is required. Using this

The data points from the curve are:

Ê

§

information and equation (6) we can find the minimum value

e

23V, I

L

e

3A, I_L7A

V

M

60V, V

B

Using the dual slope protection formulas:

0.65

of R required to prevent thermal runaway.

E

e

100 (30) (0.002)

R

0.22X

E

t

e

R

0.19X

(6)

3

E

a

30

1

e

R

2

1k

We must now use the SOA data on the TIP42, TIP41 tran-

sistors to set up the protection circuit. Below is the SOA

curve with the 4X and 8X load lines. Also shown are the

desired protection lines. Note the value of V is equal to the

B

supply voltage, so we use the formulas in the table.

b

60 0.65

0.65

e

&

91k

R

1k

1

#

J

(/2(1.8) (23)

23

e

b

&

R

3

1k

1

24k

b

7(0.22) 0.65

#

J

Note that an R of 0.22X satisfies equation (6). The final

D.C. SOA of TIP42, TIP41

Transistors

E

schematic of this amplifier is below. If the output is shorted

the current will be 1.8A and V

AC, the average power is:

is 23V. Since the input is

CE

e

&

short P

21W

D

This power is greater than was used in the heat sink calcula-

tions, so the transistors will overheat for long-duration

shorts unless a larger heat sink is used.

TL/H/7146–8

Typical Applications (Continued)

20W-8X, 30W-4X Amplifier with 1 Second Turn-ON Delay

TL/H/7146–9

j

*Additional protection for LM391N; Schottky diodes and R

100X.

9

Application Hints (Continued)

A 40W/8X, 60W/4X AMPLIFIER

Given:

Since a heat sink is required on the driver, we should inves-

tigate the output stage thermal stability at the same time to

optimize the design. If we find a value of R that is good for

E

the protection circuitry, we can then use equation (5) to find

the heat sink required for the drivers.

Power Output

40W/8X

60W/4X

Input Sensitivity

Input Impedance

1V Max

100k

The SOA characteristics of the 2N5882, 2N5880 transistors

are shown in the following curve along with a desired pro-

tection line.

g

20 Hz–20 kHz 0.25 dB

Bandwidth

Equations (1) and (2) give:

SOA 2N5882, 2N5880

e

40W/8X

60W/4X

V

25.3V

21.9V

I

3.16A

5.48A

OPeak

Therefore the supply required is:

@

30.3V 3.16A, reducing to . . .

g

@

26.9V 5.48A

g

With 15% regulation and high line we get 38.3V using

equation (3).

The minimum gain from equation (4) is:

t

A

V

18

We select a gain of 20; resulting sensitivity is 900 mV.

The input impedance and bandwidth are the same as the 20

watt amplifier so the components are the same.

e

C 5 pF

C

R

5.1k

R

100k

f

IN

1

2

TL/H/7146–10

e

10 mF

100k

C

f

The desired data points are:

The maximum supplies dictate using 80V devices. The

2N5882, 2N5880 pair are 80V, 160W transistors with a mini-

mum beta of 40 at 2A and 20 at 6A. This corresponds to a

minimum beta of 22.5 at 5.5A (I ). The MJE712,

MJE722 driver pair are 80V transistors with a minimum beta

Ê

e

I

L

e

11A

V

80V

V

B

47V

3A

I_L

M

Since the break voltage is not equal to the supply, we will

use two resistors to replace R and move V .

Opeak

3

B

Circuit Used

of 50 at 250 mA. This output combination guarantees I

with 5 mA from the LM391.

Opeak

Output transistor heat sink requirements are found using

equations (7), (9), and (10):

e

P

0.4 (60)

24W

(7)

D

b

200 55

s

e

55 C

i

6.0 C/W for T

§

(9)

§

JA

AMAX

24

s

b

e

6.0 1.1 1.0 3.9 C/W

i

(10)

§

For both output transistors on one heat sink the thermal

SA

resistance should be 1.9 C/W.

§

Now using equation (8) we find the power dissipation in the

driver:

TL/H/7146–11

24

Thevenin Equivalent

e

1.2W

P

(8)

(9)

DRIVER

20

b

150 55

s

e

i

79 C/W

§

JA

1.2

A

R₃R^B₃

e

Where: R

TH

ll

R^A₃

b

V

e

V

TH

A

R₃

B

R₃

Ð

(

a

TL/H/7146–12

10

Application Hints (Continued)

The formulas for R , R , and R do not change:

The easiest way to solve these equations is to iterate with

B

62k, then R₃

E

1

2

standard values. If we guess R^A₃

e

47.12k;

0.65

3A

e

R

E

0.22X

use 47k. The Thevenin impedance comes out 26.7k, which

is close enough to 25.55k.

b

80 0.65

e

1

e

120k

Now we will use equation (5) to determine the heat sinking

requirements of the drivers to insure thermal stability:

R

1k

R

1k

2

0.65

The formula for R now gives R when the V^ain the for-

3

TH

a

0.22 (20

1)

s

&

i

57 C/W

§

(5)

mula becomes V .

B

JA

40 (0.002)

V

B

This value is lower than we got with equation (9), so we will

use it in equation (10):

e

b

1

R

TH

2

Ê

b

w

I

_LR

E

Ð

(

s

b

e

1 50 C/W

i

57

6

(10)

§

SA

47

b

e

25.55k

1k

1

This is the required heat sink for each driver. For low TIM

we add the 1 MX resistor from pin 3 to the output and a

910k resistor from pin 4 to ground. The complete schematic

is shown below.

b

11 (0.22) 0.65

Ð

0.76 R₃

(

25.55k

V

get V .

is the additional voltage added to the supply voltage to

TH

B

a

e b

b

(V

B

e b

b

e b

17V

V

)

(47 30)

If the output is shorted, the transistor voltage is about 28V

and the current is 5A. Therefore the average power is:

TH

Now we must find R^A₃and R^B₃using the Thevenin formulas.

Putting V , V^b, and R into the appropriate formulas re-

e

70W

short PD

(/2(28) 5

TH

duces to:

TH

This is much larger than the power used to calculate the

heat sinks and the output transistors will overheat if the out-

put is shorted too long.

A

R^B₃

and

R₃R^B₃

e

ll

Typical Applications (Continued)

40W-8X, 60W-4X Amplifier

*High Frequency Ground

**Input Ground

***Speaker Ground

TL/H/7146–13

Note: All Grounds Should be Tied Together

Only at Power Supply Ground.

j

²

Additional protection for LM391N; Schottky diodes and R

100X.

11

Physical Dimensions inches (millimeters)

Molded Dual-In-Line Package (N)

Order Number LM391N-100

NS Package Number N16A

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SEMICONDUCTOR CORPORATION. As used herein:

1. Life support devices or systems are devices or

systems which, (a) are intended for surgical implant

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failure to perform, when properly used in accordance

with instructions for use provided in the labeling, can

be reasonably expected to result in a significant injury

to the user.

2. A critical component is any component of a life

support device or system whose failure to perform can

be reasonably expected to cause the failure of the life

support device or system, or to affect its safety or

effectiveness.

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Europe

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Hong Kong Ltd.

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Japan Ltd.

a

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