TPS61041DBVR全新原装原理图各脚功能电路原理芯片引脚定义-中国IC网-芯三七

TPS61040

TPS61041

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

LOW-POWER DC/DC BOOST CONVERTER IN SOT-23 AND SON PACKAGES

1

FEATURES

DESCRIPTION

•

1.8-V to 6-V Input Voltage Range

The TPS61040/41 is

a

high-frequency boost

converter dedicated for small to medium LCD bias

supply and white LED backlight supplies. The device

is ideal to generate output voltages up to 28 V from a

dual cell NiMH/NiCd or a single cell Li-Ion battery.

The part can also be used to generate standard

3.3-V/5-V to 12-V power conversions.

Adjustable Output Voltage Range up to 28 V

400-mA (TPS61040) and 250-mA (TPS61041)

Internal Switch Current

•

Up to 1-MHz Switching Frequency

28-μA Typical No-Load Quiescent Current

1-μA Typical Shutdown Current

Internal Soft Start

The TPS61040/41 operates with

a

switching

frequency up to 1 MHz. This allows the use of small

external components using ceramic as well as

tantalum output capacitors. Together with the thin

SON package, the TPS61040/41 gives a very small

overall solution size. The TPS61040 has an internal

400 mA switch current limit, while the TPS61041 has

a 250-mA switch current limit, offering lower output

voltage ripple and allows the use of a smaller form

factor inductor for lower power applications. The low

quiescent current (typically 28 μA) together with an

optimized control scheme, allows device operation at

very high efficiencies over the entire load current

range.

Available in SOT23-5 and 2 × 2 × 0.8-mm SON

Packages

APPLICATIONS

•

LCD Bias Supply

White-LED Supply for LCD Backlights

Digital Still Camera

PDAs, Organizers, and Handheld PCs

Cellular Phones

Internet Audio Player

Standard 3.3-V/5-V to 12-V Conversion

DBV PACKAGE

(Top View)

DRV PACKAGE

(Top View)

1

2

3

5

4

1

2

3

6

5

4

GND

V_IN

SW

NC

FB

V_IN

EN

SW

GND

EN

FB

TYPICAL APPLICATION

EFFICIENCY

vs

OUTPUT CURRENT

90

88

86

84

82

80

78

76

74

72

70

L1

10 µH

V

= 18 V

D1

O

V = 5 V

I

V

OUT

V

IN

to 28 V

IN

1.8 V to 6.0 V

V = 3.6 V

I

C

FF

R1

1

3

2

5

4

V

SW

IN

C

1 µF

O

V = 2.4 V

I

FB

C

4.7 µF

IN

EN

GND

R2

0.10

1

10

100

I

- Output Current - mA

O

1

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PRODUCTION DATA information is current as of publication date.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

TPS61040

TPS61041

www.ti.com

SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

(1)

ORDERING INFORMATION

SWITCH CURRENT

LIMIT, mA

PACKAGE

MARKING

T_A

PART NUMBER

PACKAGE

TPS61040DBV

TPS61041DBV

TPS61040DRV

TPS61041DRV

400

250

400

250

SOT23-5

PHOI

PHPI

CCL

–40°C to

85°C

SON-6 2×2

CAW

(1) The devices are available in tape and reel and in tubes. Add R suffix to the part number (e.g.,

TPS61040DRVR) to order quantities of 3000 parts in tape and reel or add suffix T (e.g.,

TPS61040DRVT) to order a tube with 250 pieces..

FUNCTIONAL BLOCK DIAGRAM

SW

Under Voltage

Lockout

Bias Supply

400 ns Min

VIN

FB

Off Time

Error Comparator

-

S

Power MOSFET

N-Channel

+

RS Latch

Logic

Gate

Driver

V

REF

= 1.233 V

R

Current Limit

R

SENSE

+

_

6 µs Max

On Time

EN

Soft

Start

GND

2

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

Table 1. Terminal Functions

TERMINAL

NAME DBV NO. DRV NO.

I/O

DESCRIPTION

This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown

mode reducing the supply current to less than 1 μA. This pin should not be left floating and needs

to be terminated.

EN

4

3

I

This is the feedback pin of the device. Connect this pin to the external voltage divider to program

the desired output voltage.

FB

3

4

GND

NC

2

–

1

5

–

Ground

No connection

Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to

the drain of the internal power MOSFET.

SW

V_IN

1

5

6

2

I

Supply voltage pin

DETAILED DESCRIPTION

OPERATION

The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up to

28 V. The device operates in a pulse-frequency-modulation (PFM) scheme with constant peak current control.

This control scheme maintains high efficiency over the entire load current range, and with a switching frequency

up to 1 MHz, the device enables the use of very small external components.

The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference voltage

of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon as the

inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA (TPS61041).

See the Peak Current Control section for more information. The second criteria that turns off the switch is the

maximum on-time of 6 μs (typical). This is just to limit the maximum on-time of the converter to cover for extreme

conditions. As the switch is turned off the external Schottky diode is forward biased delivering the current to the

output. The switch remains off for a minimum of 400 ns (typical), or until the feedback voltage drops below the

reference voltage again. Using this PFM peak current control scheme the converter operates in discontinuous

conduction mode (DCM) where the switching frequency depends on the output current, which results in very high

efficiency over the entire load current range. This regulation scheme is inherently stable, allowing a wider

selection range for the inductor and output capacitor.

PEAK CURRENT CONTROL

The internal switch turns on until the inductor current reaches the typical dc current limit (I_LIM) of 400 mA

(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, theactualcurrent

exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:

V

IN

I

+ I

)

100 ns

peak(typ)

LIM

L

V

IN

I

+ 400 mA )

+ 250 mA )

100 ns for the TPS61040

peak(typ)

L

V

IN

100 ns for the TPS61041

L

(1)

The higher the input voltage and the lower the inductor value, the greater the peak.

By selecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current limit

requirements. A lower current limit supports applications requiring lower output power and allows the use of an

inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output

voltage ripple as well.

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

SOFT START

All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This

can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut

down.

I

LIM

4

The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from

for 256

I

LIM

2

cycles to

for the next 256 cycles, and then full current limit (see Figure 14).

ENABLE

Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 μA (typical).

Because there is a conductive path from the input to the output through the inductor and Schottky diode, the

output voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should

not be left floating. Using a small external transistor disconnects the input from the output during shutdown as

shown in Figure 18.

UNDERVOLTAGE LOCKOUT

An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the

input voltage is below the undervoltage threshold, the main switch is turned off.

ABSOLUTE MAXIMUM RATINGS

over operating free-air temperature (unless otherwise noted)

(1)

UNIT

(2)

Supply voltages on pin V_IN

–0.3 V to 7 V

–0.3 V to V_IN+ 0.3 V

30 V

(2)

Voltages on pins EN, FB

Switch voltage on pin SW

(2)

Continuous power dissipation

Operating junction temperature

Storage temperature

See Dissipation Rating Table

–40°C to 150°C

–65°C to 150°C

260°C

T_J

T_stg

Lead temperature (soldering 10 seconds)

(1) Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings

only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating

conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) All voltage values are with respect to network ground terminal.

DISSIPATION RATING TABLE

DERATING

T

_A≤ 25°C

FACTOR

ABOVE

T_A= 70°C

POWER RATING POWER RATING

T_A= 85°C

PACKAGE

R_θJA

POWER RATING

T_A= 25°C

DBV

DRV

250°C/W

76°C/W

357 mW

3.5 mW/°C

13 mW/°C

192 mW

688 mW

140 mW

500 mW

1300 mW

4

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

RECOMMENDED OPERATING CONDITIONS

MIN

TYP

MAX UNIT

V_IN

V_OUT

L

Input voltage range

Output voltage range

Inductor⁽¹⁾

1.8

6

V

28

2.2

10

μH

MHz

μF

°C

f

Switching frequency⁽¹⁾

1

(1)

C_IN

C_OUT

T_A

Input capacitor

4.7

(1)

Output capacitor

1

–40

Operating ambient temperature

Operating junction temperature

85

T_J

125

°C

(1) See application section for further information.

ELECTRICAL CHARACTERISTICS

V_IN= 2.4 V, EN = V_IN, T_A= –40°C to 85°C, typical values are at T_A= 25°C (unless otherwise noted)

PARAMETER

SUPPLY CURRENT

TEST CONDITIONS

MIN

TYP

MAX UNIT

V_IN

Input voltage range

1.8

6

50

1

V

μA

V

I_Q

Operating quiescent current

Shutdown current

I_OUT= 0 mA, not switching, V_FB= 1.3 V

EN = GND

28

0.1

1.5

I_SD

V_UVLO

ENABLE

V_IH

Under-voltage lockout threshold

1.7

EN high level input voltage

EN low level input voltage

EN input leakage current

1.3

V

V_IL

0.4

1

I_I

EN = GND or V_IN

0.1

μA

POWER SWITCH AND CURRENT LIMIT

Vsw

t_off

Maximum switch voltage

Minimum off time

30

550

7.5

V

250

4

400

6

ns

t_on

Maximum on time

μs

R_DS(on)

MOSFET on-resistance

MOSFET leakage current

MOSFET current limit

V_IN= 2.4 V; I_SW= 200 mA; TPS61040

V_IN= 2.4 V; I_SW= 200 mA; TPS61041

V_SW= 28 V

600

750

1

1000

1250

10

mΩ

μA

mA

I_LIM

TPS61040

350

215

400

250

450

285

I_LIM

TPS61041

OUTPUT

V_OUT

V_ref

Adjustable output voltage range

Internal voltage reference

Feedback input bias current

Feedback trip point voltage

V_IN

28

1

V

1.233

I_FB

V_FB= 1.3 V

μA

V

V_FB

1.8 V ≤ V_IN≤ 6 V

1.208 1.233 1.258

1.8 V ≤ V_IN≤ 6 V; V_OUT= 18 V; I_load= 10 mA;

C_FF= not connected

(1)

Line regulation

0.05

0.15

%/V

Load regulation⁽¹⁾

V_IN= 2.4 V; V_OUT= 18 V; 0 mA ≤ I_OUT≤ 30 mA

%/mA

(1) The line and load regulation depend on the external component selection. See the application section for further information.

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

TYPICAL CHARACTERISTICS

Table 2. Table of Graphs

FIGURE

vs Load current

1, 2, 3

4

η

Efficiency

vs Input voltage

I_Q

Quiescent current

Feedback voltage

Switch current limit

vs Input voltage and temperature

vs Temperature

5

V_FB

I_SW

6

vs Temperature

7

vs Supply voltage, TPS61041

vs Supply voltage, TPS61040

vs Temperature

8

I_CL

Switch current limit

R_DS(on)

9

10

11

12

13

14

R_DS(on)

vs Supply voltage

Line transient response

Load transient response

Start-up behavior

EFFICIENCY

vs

LOAD CURRENT

vs

OUTPUT CURRENT

90

88

86

84

82

80

78

76

74

72

70

90

88

86

84

82

80

78

76

74

72

70

V

O

= 18 V

L = 10 µH

= 18 V

V

O

V = 5 V

I

TPS61040

V = 3.6 V

I

TPS61041

V = 2.4 V

I

0.10

1

10

100

0.10

1

10

100

I

O

- Output Current - mA

I - Load Current - mA

L

Figure 1.

Figure 2.

6

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

EFFICIENCY

vs

LOAD CURRENT

EFFICIENCY

vs

INPUT VOLTAGE

90

V

90

88

86

84

82

80

78

76

74

= 18 V

L = 10 µH

= 18 V

O

88

V

O

I

O

= 10 mA

86

84

82

L = 10 µH

I

O

= 5 mA

L = 3.3 µH

80

78

76

74

72

70

72

70

1

2

3

4

5

6

0.10

1

10

100

I - Load Current - mA

L

V - Input Voltage - V

I

Figure 3.

Figure 4.

TPS61040

QUIESCENT CURRENT

vs

FEEDBACK VOLTAGE

vs

FREE-AIR TEMPERATIRE

INPUT VOLTAGE

40

1.24

T

= 85°C

= 27°C

= -40°C

A

35

30

25

20

15

10

1.238

T

A

1.236

1.234

T

A

V

CC

= 2.4 V

1.232

1.23

5

0

1.8

2.4

3

3.6

4.2

4.8

5.4

6

-40 -20

0

20

40

60

80 100 120

T

A

- Temperature - °C

V - Input Voltage - V

I

Figure 5.

Figure 6.

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

TPS61040/41

SWITCH CURRENT LIMIT

vs

TPS61041

CURRENT LIMIT

vs

FREE-AIR TEMPERATURE

SUPPLY VOLTAGE

260

258

256

254

252

430

TPS61040

410

390

370

350

330

310

290

T

A

= 27°C

250

248

246

244

270

TPS61041

250

230

242

240

1.8

2.4

3

3.6

4.2

4.8

5.4

6

-40 -30 -20 -10

0

10 20 30 40 50 60 70 80 90

- Temperature - °C

T

A

V

CC

- Supply Voltage - V

Figure 7.

Figure 8.

TPS61040

TPS61040/41

CURRENT LIMIT

vs

STATIC DRAIN-SOURCE ON-STATE RESISTANCE

vs

SUPPLY VOLTAGE

FREE-AIR TEMPERATURE

420

1200

415

410

405

400

395

390

1000

800

600

400

TPS61041

T

A

= 27°C

TPS61040

200

0

385

380

−40−30 −20 −10 0 10 20 30 40 50 60 70 80 90

1.8

2.4

3

3.6

4.2

4.8

5.4

6

T

A

− Temperature − °C

V

CC

- Supply Voltage - V

Figure 9.

Figure 10.

8

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

TPS61040/41

STATIC DRAIN-SOURCE ON-STATE RESISTANCE

vs

SUPPLY VOLTAGE

1000

V

O

= 18 V

900

800

700

600

500

400

300

200

V

I

2.4 V to 3.4 V

TPS61041

TPS61040

V

O

100 mV/div

100

0

1.8

2.4

3

3.6

4.2

4.8

5.4

6

200 µS/div

Figure 12. Line Transient Response

V

− Supply Voltage − V

CC

Figure 11.

V

O

= 18 V

V

O

= 18 V

V

O

V

O

100 mA/div

5 V/div

EN

1 V/div

V

O

1 mA to 10 mA

I

50 mA/div

200 µS/div

Figure 13. Load Transient Tresponse

Figure 14. Start-Up Behavior

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APPLICATION INFORMATION

INDUCTOR SELECTION, MAXIMUM LOAD CURRENT

Because the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability

of the regulator. The selection of the inductor together with the nominal load current, input and output voltage of

the application determines the switching frequency of the converter. Depending on the application, inductor

values between 2.2 μH and 47 μH are recommended. The maximum inductor value is determined by the

maximum on time of the switch, typically 6 μs. The peak current limit of 400 mA/250 mA (typically) should be

reached within this 6-μs period for proper operation.

The inductor value determines the maximum switching frequency of the converter. Therefore, select the inductor

value that ensures the maximum switching frequency at the converter maximum load current is not exceeded.

The maximum switching frequency is calculated by the following formula:

V

(V

* V

IN(min)

OUT

L V

IN)

fS

+

max

I

P

OUT

(2)

Where:

I_P= Peak current as described in the Peak Current Control section

L = Selected inductor value

V_IN(min)= The highest switching frequency occurs at the minimum input voltage

If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step

is to calculate the switching frequency at the nominal load current using the following formula:

2 I

(V

* V ) Vd)

load

OUT

IN

_fSǒ_IloadǓ₊

2

I

L

P

(3)

Where:

I_P= Peak current as described in the Peak Current Control section

L = Selected inductor value

I_load= Nominal load current

Vd = Rectifier diode forward voltage (typically 0.3V)

A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.

The inductor value has less effect on the maximum available load current and is only of secondary order. The

best way to calculate the maximum available load current under certain operating conditions is to estimate the

expected converter efficiency at the maximum load current. This number can be taken out of the efficiency

graphs shown in Figure 1 through Figure 4. The maximum load current can then be estimated as follows:

2

I

L fS

max

* V

P

I

+ h

load max

2 (V

OUT

IN)

(4)

Where:

I_P= Peak current as described in the Peak Current Control section

L = Selected inductor value

fS_max= Maximum switching frequency as calculated previously

η = Expected converter efficiency. Typically 70% to 85%

10

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The maximum load current of the conveter is the current at the operation point where the coverter starts to enter

the continuous conduction mode. Usually the converter should always operate in discontinuous conduction

mode.

Last, the selected inductor should have a saturation current that meets the maximum peak current of the

converter (as calculated in the Peak Current Control section). Use the maximum value for I_LIMfor this calculation.

Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency

of the converter. See Table 3 and the typical applications for the inductor selection.

Table 3. Recommended Inductor for Typical LCD Bias Supply (see Figure 15)

DEVICE

INDUCTOR VALUE

10 μH

COMPONENT SUPPLIER

Sumida CR32-100

COMMENTS

High efficiency

10 μH

Sumida CDRH3D16-100

Murata LQH4C100K04

Sumida CDRH3D16-4R7

Murata LQH3C4R7M24

High efficiency

TPS61040

10 μH

High efficiency

4.7 μH

Small solution size

4.7 μH

High efficiency

Small solution size

TPS61041

10 μH

Murata LQH3C100K24

SETTING THE OUTPUT VOLTAGE

The output voltage is calculated as:

R1

R2

_{+ 1.233 V}ǒ_{1 )}Ǔ

V

OUT

(5)

For battery-powered applications, a high-impedance voltage divider should be used with a typical value for R2 of

≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity of

the feedback pin.

A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the

error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS61040/41 shows

double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage

ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.

The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good

starting point is to use a 10-pF feedforward capacitor. As a first estimation, the required value for the feedforward

capacitor at the operation point can also be calculated using the following formula:

1

C

+

FF

fS

20

2 p

R1

(6)

Where:

R1 = Upper resistor of voltage divider

fS = Switching frequency of the converter at the nominal load current (See the INDUCTOR SELECTION,

MAXIMUM LOAD CURRENT section for calculating the switching frequency)

C_FF= Choose a value that comes closest to the result of the calculation

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The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for line

regulation is paramount, the selected feedforward capacitor should be as small as possible. See the LINE AND

LOAD REGULATION section for more information about line and load regulation.

LINE AND LOAD REGULATION

The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV

peak-to-peak voltage ripple on the feedback pin FB gives good results.

Some applications require a very tight line regulation and can only allow a small change in output voltage over a

certain input voltage range. If no feedforward capacitor C_FFis used across the upper resistor of the voltage

feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage

ripple is higher because the TPS61040/41 shows output voltage bursts instead of single pulses on the switch pin

(SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage

ripple.

If a larger output capacitor value is not an option, a feedforward capacitor C_FFcan be used as described in the

previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the

feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.

There are two ways to improve the line regulation further:

1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,

as well as the voltage ripple on the feedback pin.

2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback pin

down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward capacitor

C_FFcan be used.

OUTPUT CAPACITOR SELECTION

For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low

ESR value but tantalum capacitors can be used as well, depending on the application.

Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output

voltage ripple can be calculated as:

I

L

I

out

1

P

DV

+

–

) I ESR

ǒ

Ǔ

out

P

C

fS(Iout) Vout ) Vd–Vin

out

(7)

where:

I_P= Peak current as described in the Peak Current Control section

L = Selected inductor value

I_out= Nominal load current

fS (I_out) = Switching frequency at the nominal load current as calculated previously

Vd = Rectifier diode forward voltage (typically 0.3 V)

C_out= Selected output capacitor

ESR = Output capacitor ESR value

See Table 4 and the typical applications section for choosing the output capacitor.

Table 4. Recommended Input and Output Capacitors

DEVICE

CAPACITOR

4.7 μF/X5R/0805

10 μF/X5R/0805

1 μF/X7R/1206

1 μF/X5R/1206

4.7 μF/X5R/1210

VOLTAGE RATING

COMPONENT SUPPLIER

Tayo Yuden JMK212BY475MG

Tayo Yuden JMK212BJ106MG

Tayo Yuden TMK316BJ105KL

Tayo Yuden GMK316BJ105KL

Tayo Yuden TMK325BJ475MG

COMMENTS

C_IN/C_OUT

C_OUT

6.3 V

25 V

35 V

25 V

TPS61040/41

C_OUT

12

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

INPUT CAPACITOR SELECTION

For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 μF ceramic input capacitor

is sufficient for most of the applications. For better input voltage filtering this value can be increased. See Table 4

and typical applications for input capacitor recommendations.

DIODE SELECTION

To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the

peak current rating of the converter as it is calculated in the Peak Current Control section. Use the maximum

value for I_LIMfor this calculation. See Table 5 and the typical applications for the selection of the Schottky diode.

Table 5. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15)

DEVICE

REVERSE VOLTAGE

COMPONENT SUPPLIER

ON Semiconductor MBR0530

ON Semiconductor MBR0520

ON Semiconductor MBRM120L

Toshiba CRS02

COMMENTS

30 V

20 V

30 V

TPS61040/41

High efficiency

LAYOUT CONSIDERATIONS

Typical for all switching power supplies, the layout is an important step in the design; especially at high peak

currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems

and duty cycle jitter.

The input capacitor should be placed as close as possible to the input pin for good input voltage filtering. The

inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into

other circuits. Because the feedback pin and network is a high-impedance circuit, the feedback network should

be routed away from the inductor. The feedback pin and feedback network should be shielded with a ground

plane or trace to minimize noise coupling into this circuit.

Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground

plane minimizes ground shifts and noise.

D1

L1

V

O

C

FF

R1

V

IN

SW

FB

IN

C

O

C

IN

R2

EN

GND

Figure 15. Layout Diagram

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13

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

L1

10 µH

D1

V

18 V

OUT

V

IN

1.8 V to 6 V

TPS61040

C

22 pF

FF

R1

2.2 MW

V

IN

SW

FB

C2

1 µF

C1

4.7 µF

L1:

D1:

C1:

C2:

Sumida CR32-100

Motorola MBR0530

Tayo Yuden JMK212BY475MG

Tayo Yuden TMK316BJ105KL

EN

GND

R2

160 kW

Figure 16. LCD Bias Supply

L1

10 µH

D1

V

O

18 V

TPS61040

C

FF

R1

2.2 MW

22 pF

V

IN

SW

FB

IN

C2

1 µF

1.8 V to 6 V

C1

4.7 µF

DAC or Analog Voltage

0 V = 25 V

1.233 V = 18 V

EN

GND

R2

160 kW

L1:

Sumida CR32-100

D1:

C1:

C2:

Motorola MBR0530

Tayo Yuden JMK212BY475MG

Tayo Yuden GMK316BJ105KL

Figure 17. LCD Bias Supply With Adjustable Output Voltage

R3

200 kW

BC857C

L1

10 µH

D1

V

IN

V

OUT

1.8 V to 6 V

18 V / 10 mA

TPS61040

R1

C

FF

2.2 MW

22 pF

V

IN

SW

FB

C2

1 µF

C3

0.1 µF

(Optional)

C1

4.7 µF

R2

160 kW

EN

GND

L1:

D1:

C1:

C2:

Sumida CR32-100

Motorola MBR0530

Tayo Yuden JMK212BY475MG

Tayo Yuden TMK316BJ105KL

Figure 18. LCD Bias Supply With Load Disconnect

14

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TPS61041

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

D3

V2 = -10 V/15 mA

D2

C4

4.7 µF

C3

1 µF

L1

D1

6.8 µH

V1 = 10 V/15 mA

TPS61040

C

FF

22 pF

R1

1.5 MW

V

SW

FB

IN

V

IN

= 2.7 V to 5 V

C2

1 µF

C1

4.7 µF

L1:

Murata LQH4C6R8M04

D1, D2, D3: Motorola MBR0530

EN

GND

R2

C1:

Tayo Yuden JMK212BY475MG

210 kW

C2, C3, C4: Tayo Yuden EMK316BJ105KF

Figure 19. Positive and Negative Output LCD Bias Supply

L1

6.8 µH

D1

V

O =

12 V/35 mA

TPS61040

C

4.7 pF

FF

R1

1.8 MW

V

IN

3.3 V

V

IN

SW

FB

C2

4.7 µF

C1

10 µF

L1:

Murata LQH4C6R8M04

Motorola MBR0530

Tayo Yuden JMK212BJ106MG

Tayo Yuden EMK316BJ475ML

EN

GND

R2

205 kW

D1:

C1:

C2:

Figure 20. Standard 3.3-V to 12-V Supply

D1

3.3 µH

TPS61040

5 V/45 mA

C

FF

3.3 pF

R1

620 kW

V

1.8 V to 4 V

SW

FB

IN

C2

4.7 µF

C1

4.7 µF

R2

200 kW

EN

GND

L1:

D1:

Murata LQH4C3R3M04

Motorola MBR0530

C1, C2: Tayo Yuden JMK212BY475MG

Figure 21. Dual Battery Cell to 5-V/50-mA Conversion

Efficiency Approx. Equals 84% at V_IN= 2.4 V to Vo = 5 V/45 mA

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SLVS413C–OCTOBER 2002–REVISED AUGUST 2007

L1

D1

10 µH

D2

24 V

(Optional)

V

CC

= 2.7 V to 6 V

V

IN

SW

FB

C1

4.7 µF

L1:

D1:

C1:

C2:

Murata LQH4C100K04

Motorola MBR0530

Tayo Yuden JMK212BY475MG

Tayo Yuden TMK316BJ105KL

C2

1 µF

R

82 Ω

EN

PWM

100 Hz to 500 Hz

GND

S

Figure 22. White LED Supply With Adjustable Brightness Control

Using a PWM Signal on the Enable Pin, Efficiency Approx. Equals 86% at V_IN= 3 V, I_LED= 15 mA

D1

L1

MBRM120L

10 µH

†

C2

D2

24 V

(Optional)

V

CC

= 2.7 V to 6 V

100 nF

V

SW

FB

IN

C1

4.7 µF

R1

120 kW

EN

GND

R

S

110 W

L1:

D1:

C1:

C2:

Murata LQH4C3R3M04

Motorola MBR0530

Tayo Yuden JMK212BY475MG

Standard Ceramic Capacitor

Analog Brightness Control

3.3 V Led Off

R2 160 kW

0 V Iled = 20 mA

A. A smaller output capacitor value for C2 causes a larger LED ripple.

Figure 23. White LED Supply With Adjustable Brightness Control

Using an Analog Signal on the Feedback Pin

16

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Product Folder Link(s): TPS61040 TPS61041

PACKAGE OPTION ADDENDUM

www.ti.com

15-Oct-2007

PACKAGING INFORMATION

Orderable Device

TPS61040DBVR

Status ⁽¹⁾

ACTIVE

Package Package

Pins Package Eco Plan ⁽²⁾Lead/Ball Finish MSL Peak Temp ⁽³⁾

Qty

Type

Drawing

SOT-23

DBV

5

3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM

no Sb/Br)

TPS61040DBVRG4

SOT-23

DBV

5

3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM

no Sb/Br)

TPS61040DRVR

TPS61041DBVR

PREVIEW

ACTIVE

SON

DRV

DBV

6

5

3000

TBD

Call TI

SOT-23

3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM

no Sb/Br)

TPS61041DBVRG4

ACTIVE

SOT-23

DBV

5

3000 Green (RoHS & CU NIPDAU Level-1-260C-UNLIM

no Sb/Br)

TPS61041DRVR

TPS61041DRVT

PREVIEW

SON

DRV

5

3000

250

TBD

Call TI

⁽¹⁾The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in

a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2)

Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check

http://www.ti.com/productcontent for the latest availability information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements

for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered

at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and

package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS

compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame

retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material)

(3)

MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder

temperature.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is

provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the

accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take

reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on

incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited

information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI

to Customer on an annual basis.

Addendum-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

5-Oct-2007

TAPE AND REEL BOX INFORMATION

Device

Package Pins

Site

Reel

A0 (mm)

B0 (mm)

K0 (mm)

P1

W

Pin1

Diameter Width

(mm) (mm) Quadrant

(mm)

179

(mm)

TPS61040DBVR

TPS61041DBVR

DBV

5

SITE 48

8

3.2

1.4

4

8

Q3

179

Pack Materials-Page 1

PACKAGE MATERIALS INFORMATION

www.ti.com

5-Oct-2007

Device

Package

Pins

Site

Length (mm) Width (mm) Height (mm)

TPS61040DBVR

TPS61041DBVR

DBV

5

SITE 48

195.0

200.0

45.0

Pack Materials-Page 2

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