AD633
the basis of voltage controlled integrators and oscillators as will
be shown later in this Applications section. The transfer func-
tion of this circuit has the form
I
O
=
1
(
X
R
1
dB
f2 f1
+15V
0.1 F
0
–6dB/OCTAVE
OUTPUTA
OUTPUT B =
f
−
X
2
Y
1
−
Y
2
10
V
)
(
)
(Equation 7)
CONTROL
INPUT E
C
SIGNAL
INPUT E
S
1
2
3
4
X1
X2
Y1
Y2
+V
S 8
W
7
Z
6
–V
S 5
0.1 F
–15V
OUTPUTB
AD633JN
Linear Amplitude Modulator
The AD633 can be used as a linear amplitude modulator with
no external components. Figure 10 shows the circuit. The car-
rier and modulation inputs to the AD633 are multiplied to
produce a double-sideband signal. The carrier signal is fed
forward to the AD633’s Z input where it is summed with the
double-sideband signal to produce a double-sideband with carrier
output.
Voltage Controlled Low-Pass and High-Pass Filters
1 + T
1
P
1 + T
2
P
R
1
OUTPUT A =
1 + T
2
P
C
T
1
= 1 = RC
W
1
T
2
=
10
1
=
W
2
E
C
R
C
Figure 11. Voltage Controlled Low-Pass Filter
dB
f1 f2
+15V
0.1 F
CONTROL
INPUT E
C
SIGNAL
INPUT E
S
1
2
3
4
0
f
Figure 11 shows a single multiplier used to build a voltage con-
trolled low-pass filter. The voltage at output A is a result of
filtering, E
S
. The break frequency is modulated by E
C
, the con-
trol input. The break frequency, f
2
, equals
f
2
=
X1
X2
Y1
Y2
+V
S 8
W
7
OUTPUTB
+6dB/OCTAVE
OUTPUTA
OUTPUT B
C
OUTPUT A
R
AD633JN
Z
6
–V
S 5
0.1 F
–15V
(
E
C
20
V
π
RC
)
(Equation 8)
and the rolloff is 6 dB per octave. This output, which is at a
high impedance point, may need to be buffered.
The voltage at output B, the direct output of the AD633, has
same response up to frequency f
1
, the natural breakpoint of RC
filter,
f
1
=
1
2
π
RC
Figure 12. Voltage Controlled High-Pass Filter
Voltage Controlled Quadrature Oscillator
(Equation 9)
then levels off to a constant attenuation of f
1
/f
2
= E
C
/10.
+15V
0.1 F
MODULATION
INPUT
E
M
CARRIER
INPUT
E
C
sin t
1
2
3
4
X1
X2
Y1
Y2
+V
S 8
W
7
Z
6
–V
S 5
0.1 F
–15V
W = 1+
E
M
10V
E
C
sin
t
AD633JN
Figure 13 shows two multipliers being used to form integrators
with controllable time constants in a 2nd order differential
equation feedback loop. R2 and R5 provide controlled current
output operation. The currents are integrated in capacitors C1
and C2, and the resulting voltages at high impedance are applied
to the X inputs of the “next” AD633. The frequency control
input, E
C
, connected to the Y inputs, varies the integrator gains
with a calibration of 100 Hz/V. The accuracy is limited by the
Y-input offsets. The practical tuning range of this circuit is
100:1. C2 (proportional to C1 and C3), R3, and R4 provide
regenerative feedback to start and maintain oscillation. The
diode bridge, D1 through D4 (1N914s), and Zener diode D5
provide economical temperature stabilization and amplitude
stabilization at
±
8.5 V by degenerative damping. The out-
put from the second integrator (10 V sin
ωt)
has the lowest
distortion.
AGC AMPLIFIERS
Figure 10. Linear Amplitude Modulator
For example, if R = 8 kΩ and C = 0.002
µF,
then output A has
a pole at frequencies from 100 Hz to 10 kHz for E
C
ranging
from 100 mV to 10 V. Output B has an additional zero at 10 kHz
(and can be loaded because it is the multiplier’s low impedance
output). The circuit can be changed to a high-pass filter Z inter-
changing the resistor and capacitor as shown in Figure 12 below.
Figure 14 shows an AGC circuit that uses an rms-dc converter
to measure the amplitude of the output waveform. The AD633
and A1, 1/2 of an AD712 dual op amp, form a voltage con-
trolled amplifier. The rms dc converter, an AD736, measures
the rms value of the output signal. Its output drives A2, an
integrator/comparator, whose output controls the gain of the
voltage controlled amplifier. The 1N4148 diode prevents the
output of A2 from going negative. R8, a 50 kΩ variable resistor,
sets the circuit’s output level. Feedback around the loop forces
the voltages at the inverting and noninverting inputs of A2 to be
equal, thus the AGC.
REV. B
–5–
AD [ ANALOG DEVICES ]
